More specifically, the invention finds an application in fixed or mobile digital television receivers, complying specifically with the DVB standard, for filtering interfering signals such as, for example, WIMAX signals (Worldwide Interoperability for Microwave Access), present in the frequency band known as digital dividend band.
The digital dividend represents the frequency resources that will be liberated by the passage of television broadcasting from the analog mode to the digital mode. The digital dividend band varies from one region of the world to another and is comprised, for instance, between 698 MHz and 862 MHz for the Americas and 790 MHz and 862 MHz for Europe/Asia as shown in FIG. 1.
These liberated frequency bands will be dedicated to both mobile digital television broadcasting and telecommunications applications. They are particularly sought after by telecommunications operators and broadcasters, due to a superior level of efficiency with respect to frequencies higher than 1 GHz, in terms of coverage and penetration of buildings, and in terms of very much lower costs for the creation and operation of networks. They can, for instance, be used for transmitting new signals such as WIMAX signals. These new signals then constitute an interference source for the reception of DVB signals. Moreover, when the DVB receiver and the WIMAX transmitter are present in a same terminal (multimode and multistandard terminal), the WIMAX signals can saturate the DVB receiver. It is therefore deemed necessary to filter these interfering signals before processing the DVB signals. The WIMAX signals must be filtered in bands that vary according to region. Nevertheless, it is known that the signal must be attenuated in a 10 MHz wide band and that the WIMAX signal must be rejected by 42 dB. There thus exists certain cases when the WIMAX transmitter will not be present and where a filter will not be useful.
The filtering of these interfering signals can be achieved using an appropriate band-rejection or stop band filter. It is thus known in the art to use a half-wave in-line resonator filter such as shown in FIG. 2, which is dimensioned to resonate at a high frequency, well above the useful frequency. The filter of FIG. 2 comprises a transmission line LT (Zo, Θ, k) to which a λ/2 resonator RE is coupled. At resonance frequency level, the energy from the transmission line is “absorbed” by the resonator, thus creating a theoretically infinite attenuation in a relatively narrow band around the resonance frequency. Nevertheless, the disadvantages in using this filter consist in the production of losses that considerably deteriorate the rejection. It is also more cumbersome and difficult to tune on the central frequency. In order to resolve the problem of size, it has been proposed to connect a capacitor C1 to one end of the resonator RE, as shown in FIG. 3. In this case, its equivalent electrical length increases and the frequency of the rejected band decreases. In order for the resonator to reject a particular frequency one must choose the capacitor value suitably. Nevertheless, this capacitor C1 is not ideal, as it has a parasitic resistance Rs that increases when the value of the capacitor increases. This resistance can, under certain conditions, degrade the global quality factor of the loaded line. This is all the more critical since, in order to assign this type of structure, resorting to varactors which display heavy losses is unavoidable.
In order to solve the above problems, it has been proposed to replace the load capacitor by a negative resistance circuit simulating an active capacitor such as represented in FIG. 4, the left part representing the negative resistance circuit and the right part the parallel equivalent model. On the part of FIG. 4, the negative resistance circuit is constituted of a transistor Q1 having base connected to a resonator RE and collector connected to a serial LCR circuit. As shown on the right side of FIG. 4, this circuit is the equivalent of a resistor Rneg parallel to a capacitor Ceq. This circuit enables easy adjustment of the load capacitor value using a single bias voltage while ensuring the compensation of the filter losses, which ensures a high quality factor. The simulation results are shown in FIG. 5 which gives, as a function of the frequency, the transmission (curve a) and the rejection (curve b) of the filter. These curves show a very high rejection (>42 dB) at the central filter frequency around 700 MHz. The latter can be tuned by a simple adjustment of the bias voltage.